System and method for orthogonal frequency division multiplexed optical communication

ABSTRACT

A system for optical communication send optical signals over a plurality of wavelength channels. Each wavelength channel comprises a number of orthogonal subchannel frequencies which are spaced apart from one another by a predetermined amount. Each of the subchannel frequencies is modulated with data from a data stream. The data modulation scheme splits a subchannel frequency code into H and V components, and further processes the components prior to modulation with data. The various data-modulated subchannels are then combined into a single channel for transmission. The received signals are detected and demodulated with the help of a symbol timing recovery module which establishes the beginning and end of each symbol. A polarization mode distortion compensation module at the receiver is used to mitigate the effects to polarization more distortion in the fiber.

RELATED APPLICATIONS

The present application is a continuation-in-part of U.S. applicationSer. No. 09/962,243, filed Sep. 26, 2001, which itself claims priorityto U.S. provisional application No. 60/234,930, filed Sep. 26, 2000.

FIELD OF THE INVENTION

The present invention relates to the field of optical communicationsystems utilizing modulation techniques to obtain high spectralefficiency.

BACKGROUND OF THE INVENTION

Dense wavelength division multiplexing (DWDM) increases the capacity ofembedded fiber by assigning incoming optical signals to specificfrequencies (wavelength, lambda) within a designated frequency band andthen multiplexing the resulting signals out onto one fiber. DWDMcombines multiple optical signals so that they can be amplified as agroup and transported over a single fiber to increase capacity of thetelecommunication network. Each signal carried can be at a differentrate (OC-3/12/24, etc.) and in a different format (SONET, ATM, data,etc.). Limiting bandwidth of the useable band of the optical fiber toaccommodate future growth is the driving force behind the effort toincrease the spectral efficiency of DWDM systems.

FIG. 1 is a block diagram of a prior art simplex DWDM system. A DWDMmultiplexer 110 combines several optical signals, hereinafter referredto as channels, into a single multi-channel optical signal that istransmitted through the optical fiber 120. Optical amplifiers 125 may beconnected to the optical fiber 120 to amplify the optical signal.Conversely, the DWDM demultiplexer 130 receives the multi-channeloptical signal transmitted through the optical fiber 120 and splits itinto separate channels. Each channel is characterized by a distinctwavelength designated as λ_(i) in FIG. 1 where the index, i, runs from 1to N where N is the number of channels in the DWDM system. It isunderstood that a wavelength has a corresponding frequency f_(i) andthat one may refer to either frequency or wavelength while meaning thesame physical attribute of the signal. For an N-channel DWDM system,there are N transmitters 140 and N receivers 150 with one transmitter140 and one receiver 150 for each channel. A transmitter 140 generatesthe optical carrier signal at the channel wavelength and modulates thecarrier signal with a single data stream before transmitting themodulated optical signal to the multiplexer 110. The multiplexer 110then combines the N modulated optical signals having different channelwavelengths into a single multi-channel optical signal, and sends thisthrough the fiber 120. The demultiplexer 130 receives the multi-channeloptical signal and separates it into the different channel wavelengths.Each receiver 150 then demodulates one of the demultiplexed channelsignals to extract the data signal. While FIG. 1 shows a prior artsimple system, it is understood that in real life, a duplex system isused, with one or more transmitters and receivers at each end. Dutton,Harry J. R., Understanding Optical Communications, 1998, pp. 513–568,ISBN 0-13-020141-3 presents a description of the DWDM system and of itscomponents and is herein incorporated by reference.

The data rate (in bits per second or bps) through a single optical fibermay be increased by combining one or more of the following methods:increasing the data modulation rate; increasing the number of channelsper fiber; and selecting a modulation method having a higher spectralefficiency.

Increasing the data modulation rate is limited by semiconductortechnology and cost, as well as frequency-dependent fiber impairments aschromatic and Polarization Mode Dispersion (PMD). Increasing the numberof channels per fiber is limited by the properties of optical componentmaterials. Current and proposed implementations of DWDM systems use achannel modulation rate of about 10 GHz (OC-192) and use 40 channelsover the conventional optical band (C-band) between 1530 nm and 1560 nm.Therefore, the transmission bit-rate through a single optical fiber isabout 400 Gbps. Each channel has a bandwidth of about 100 GHz. Thespectral efficiency is defined as the channel bit-rate divided by thechannel bandwidth. The spectral efficiency of the system is therefore0.1 bit/Hz. The spectral efficiency may be doubled by using a coherentmodulation technique such as quadrature phase shift keying (QPSK). QPSKencodes two bits per modulation period and therefore doubles the channeltransmission bit-rate to 20 Gbps. The two bits encoded during QPSK arereferred to as a symbol and the modulation period is referred to as thesymbol period. The inverse of the symbol period is the symbol rate.

The channel bit-rate may also be doubled by combining two data streamsinto a single channel. U.S. Pat. No. 6,038,357 issued to Pan discloses afiber optic network that combines two data streams into a single channelby polarizing the optical signal modulated by the first data stream to apolarization plane that is orthogonal to the polarization plane of theoptical signal modulated by the second data stream.

Polarization mode dispersion (PMD) arises in optical fiber when circularsymmetry is broken by the presence of an elliptical core or bynoncircularly symmetric stresses. The loss of circular symmetry resultsin the difference in the group velocities associated with the twopolarization modes of the fiber. The main effect of the PMD is thesplitting of the narrow-band pulse into two orthogonally polarizedpulses (dual imaging) that propagate through the fiber with thedifferent group velocities. As the dual images propagate through thebirefringent fiber, there states of polarization (SOP) constantlyundergo changes causing the random coupling between the two images.

The PMD varies randomly from fiber to fiber. In the single fiber, thePMD also varies randomly with the optical carrier frequency and ambienttemperature. PMD broadens and degrades the signal and limits thedistance the signal may propagate before the information encoded in thesignal is lost.

Therefore, there remains a need to improve the spectral efficiency ofexisting/planned DWDM standards (OC-48 at 2.048 Gbps or OC-192 at 10Gbps) using existing fiber optic cables. There also remains a need forPMD compensation of the received optical signal.

SUMMARY OF THE INVENTION

In one aspect, the present invention is directed to a subchannelfrequency division multiplexing (FDM) system comprising a transmitterand receiver. The transmitter allows each DWDM channel to carry Ksubchannels, each subchannel modulated with a separate data stream. Eachsubchannel is characterized by a subchannel frequency, f_(k), and eachsubchannel frequency is separated from adjacent subchannels by aconstant frequency spacing, Δf. The receiver extracts each subchannel bymixing the channel signal with a local reference laser having the samefrequency as the subchannel frequency. The symbol information isrecovered by integrating the mixed signal over a symbol period 1/Δf, ora part thereof.

In another aspect, the present invention is directed to an opticalcommunication transmitter. The transmitter includes at least one lightsource arranged to output a number K subchannel light beams, eachsubchannel light beam being spaced apart from adjacent subchannel lightbeams by a constant subchannel frequency spacing Δf. The transmitteralso includes K data modulators, one to modulate each subchannel lightbeam with data from a data stream, wherein each data modulator includesa polarization beam combiner which encodes data from said data stream ontwo orthogonal polarizations to form a subchannel signal. Thetransmitter also may include a combiner configured to the combine Ksubchannel signals to thereby form an optical channel signal.

The transmitter may include a pulse shaper circuit to shape eachsubchannel light beam prior to modulation with the data.

The transmitter's light source may comprise individual light sources,each outputting a single frequency. A frequency calibration circuit maybe used to ensure even amplitude and spacing between the outputs of theindividual light sources. Frequency calibration of the subchannelcarrier signals is accomplished by determining the mean signal of thesubchannel carrier mixed with a delayed version of itself. The frequencyseparation, Δω is proportional to the difference between the meansignals of the two subchannel carriers. Alternatively, the light sourcemay comprise a single frequency comb generator that outputs a signalhaving evenly spaced, even-amplitude frequencies. Regardless of the typeof light source used, the receiver may employ a frequency offsetcompensator.

In yet another aspect, the present invention is directed to an opticalcommunication homodyne receiver that uses coherent detection. Thehomodyne receiver includes an optical splitter that output multipleidentical copies of an incoming signal, a number K subchannel receivers,the k^(th) subchannel receiver including optical and digital circuitryconfigured to receive the k^(th) of said K identical received channelsignals and a reference light beam having a subchannel frequency f_(k),and output a first digital signal representative of in-phase andquadrature components of a first orthogonal polarization componentassociated with subchannel frequency f_(k), and also output a seconddigital signal representative of in-phase and quadrature components of asecond orthogonal polarization component associated with subchannelfrequency f_(k), the first and second digital signals containinginformation representative of the data stream that was used to modulatethe k^(th) subchannel signal, and a receiver processor configured toreceive said first and second digital signals and output one of said Kdata streams.

A frequency calibration system for calibrating a number K of laser lightbeams may be used in conjunction with either the transmitter or ahomodyne receiver. The frequency calibration system includes an opticalswitch system configured to select one from among the K laser lightbeams and a reference beam and output a selected beam, a splitterdisposed to receive the selected beam and output first identical firstand second selected beams, an optical detector configured to receive adelayed version of the first selected beam and the second selected beam,and output at least one electrical signal proportional to a phasedifference between the two beams, and a controller configured to receivesaid at least one electrical signal and output at least one frequencycalibration control signal to control at least one light sourceresponsible for creating at least one of said plurality of laser lightbeams.

In yet another aspect, the present invention is directed to an opticalcommunication self-homodyne receiver that uses coherent detection. Theself-homodyne receiver extracts a phase difference between an incomingsignal, and a one-symbol delayed version of that signal, thus obviatingthe need for a reference light beam. The phase difference may correspondto a differential quadrature phase shift keying (DQPSK) encoding schemefor the data.

For both the homodyne and the self-homodyne receivers, the receiverprocessor may include polarization mode dispersion (PMD) compensationcontrol module, a synchronization and symbol timing module, and a datademodulation module.

The PMD compensation control module (PCM) associated with the receivercalculates the rotations needed to correct for Polarization ModeDistortion (PMD) experienced by an optical signal as it is sent from atransmitter to a receiver. The PCM optimizes the signals representingthe received symbol by applying a rotation transformation to thesignals.

A polarization mode dispersion (PMD) compensator device may be employedwith either homodyne or a self-homodyne receiver. In such case, theangles calculated by the PCM are used to drive an optical polarizationcompensator device to correct for PMD. The PMD compensator device may beadjusted as each pair of candidate rotation angles is attempted duringthe search for an optimum pair of such angles. A metric, such as anenvelope stability metric, may be used to help determine the optimumpair of rotation angles.

Instead of a PMD compensator device, the receiver's PMD compensationcontrol module may provide for a fully digitally-implemented(“all-digital”) polarization compensation. In such an embodiment,polarization compensation is done entirely digitally without the use ofan optical polarization compensator device. The matrix componentscomprising the rotation transformation are updated in parallel to thesignal processing of the received symbols thereby eliminating artificialdegradations to the signals caused by the PCM's search for the optimumrotation. The optimum rotation is determined by a gradient search basedon criteria matched to the data format. The digital PCM eliminates theneed for an optical polarization compensator thereby significantlyreducing the cost of the optical communication receiver. A metric, suchas an envelope stability metric, may be used to help determine theoptimum pair of rotation angles for the rotation transformation.

The symbol timing module may employ a discriminator function based onthe Stokes parameters of a received subchannel signal. The symbol timingmodule may also employ a Muller and Muller symbol timing recoveryscheme. In the case of the “all-digital” polarization compensation, ahybrid symbol timing module employing both a Stokes parameter-basedscheme and a Muller and Muller based scheme may be employed. The Stokesparameter based scheme calculates a discriminator function based oninner products of Stokes vectors derived from first and second digitalsignals representing first and second polarization components of anoptical signal.

BRIEF DESCRIPTION OF THE FIGURES

The present invention may be understood by reference to the followingdetailed description of the preferred embodiment of the presentinvention, illustrative examples of specific embodiments of theinvention and the appended figures in which:

FIG. 1 is a block diagram of a simplex prior art DWDM systemarchitecture;

FIG. 2 a is a block diagram of a simplex optical communication system inaccordance with a preferred embodiment of the present invention;

FIG. 2 b is a block diagram of one-half of a duplex opticalcommunication system in accordance with a preferred embodiment of thepresent invention;

FIG. 3 a is a block diagram of a first embodiment of a transmitter inaccordance with the present invention.

FIG. 3 b is a block diagram of a second embodiment of a transmitter inaccordance with the present invention.

FIG. 4 is a block diagram of the data modulator used in the transmittershown in FIG. 3 a & 3 b.

FIG. 5 a is a block diagram of a one embodiment of the receiver shown inFIGS. 2 a and 2 b.

FIG. 5 b is a block diagram of another embodiment of the receiver shownin FIGS. 2 a and 2 b.

FIG. 5 c is a functional diagram of the receiver processor seen in FIG.5 a.

FIG. 5 d is a functional diagram of the receiver processor seen in FIG.5 b.

FIG. 6 is a block diagram of the frequency calibrator in a preferredembodiment of the present invention.

FIG. 7 is a block diagram of the optical phase detector in a preferredembodiment of the present invention.

FIG. 8 is a Poincare' sphere representing the polarization state of asignal.

FIG. 9 is a flow diagram of the PMD controller.

FIG. 10 is a graph of the discriminator function used in the Stokes'based timing error detector.

FIG. 11 a shows a first embodiment of a self-homodyne opticalcommunication receiver in accordance with the present invention.

FIG. 11 b shows a second embodiment of a self-homodyne opticalcommunication receiver in accordance with the present invention.

FIG. 11 c shows a third embodiment of a self-homodyne opticalcommunication receiver in accordance with the present invention.

FIG. 12 shows an optical hybrid network circuit which receive two inputsignals and outputs four combined complex signals.

FIG. 13 shows a detection circuit which receives two analog inputsignals representing in-phase and quadrature components and outputs twocorresponding digital signals.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 2 a is a block diagram of the system architecture of a simplex DWDMcommunication system 249 in accordance with the present invention. Thecommunication system 249 has a transmitter end 259 provided withmultiple transmitters 250 and a receiver end 269 provided with multiplereceivers 260.

Each transmitter 250 outputs an optical signal, hereinafter referred toas a channel. Each channel output by a transmitter 250 is characterizedby a family or group of distinct wavelengths designated as {λ_(i)} inFIG. 2 a where the index, i, runs from 1 to N where N is the number ofchannels in the DWDM system. For an N-channel DWDM system, there are Ntransmitters 250 and N receivers 260 with one transmitter 250 and onereceiver 260 for each channel.

In the communication system 249, a DWDM multiplexer 210 combines thechannels output by the transmitters into a composite optical signal {λ}for transmission through the optical fiber 220. Optical amplifiers 222may be connected to the optical fiber 220 to amplify the optical signal.At the receiver end 269, the DWDM demultiplexer 230 receives thecomposite optical signal {λ} transmitted through the optical fiber 220and splits the received composite optical signal into the separatechannels {λ_(i)} for further processing.

Each channel transmitter 250 combines several optical signals,hereinafter referred to as subchannels, into a single channel signal.Each subchannel is characterized by a wavelength, λ_(ik), where theindex i identifies the channel and the index k identifies thesubchannel. Each subchannel is capable of carrying a data stream. Thus,each channel transmitter is capable of supporting a total of Ksubchannels to thereby transmit K data streams. Conversely, the channelreceiver 260 demodulates the channel signal to extract the K datasignals 290.

As seen in FIG. 2 a, each transmitter 250 receives, for its channel, atotal of K data streams, the data streams being depicted by DATA_(ijk),where subscript ‘i’ is the channel index, j is the polarization index(either ‘H’ or ‘V’) and subscript ‘k’ is the data stream index. For agiven ‘i’ and ‘k’, the ‘H’ and ‘V’ data streams are preferably used tomodulate the same subchannel on H and V polarizations. Thus, each of Ksubchannels, when transmitted, is used on both polarizations, and eachpolarization carries different data DATA_(iHk) and DATA_(iVk). Thus,since the index ‘j’ takes on only two possible values, one may considerthe incoming data streams to effectively comprise 2K data streams, eachhaving a first component that will be sent on H polarization on somesubchannel λ_(ik) and a second component that will be sent on Vpolarization on the same subchannel λ_(ik).

While the above ‘data stream’ nomenclature may be expedient toillustrate the use of both H and V polarization in accordance with thepresent invention, it should be kept in mind that a single data streammay be formatted in any number of ways including the “DATA_(ijk)”representation discussed above. What is important is that the incomingdata is used to modulate each subchannel on both H and V polarizations,with a total of K subchannels being combined and transmitted on eachchannel. Each subchannel includes both the ‘H’ and V′ data components,i.e., both ‘j’ values from a “DATA_(ijk)” data stream and these are senton the same frequency.

FIG. 2 b is a block diagram of the system architecture of anotherembodiment of the present invention showing one end of a bi-directionalcommunication link. A DWDM multiplexer (MUX) 210 is connected to adownstream optical fiber 220. Similarly, a DWDM demultiplexer (DMUX) 230is connected to an upstream optical fiber 225. At the other end (notshown) of the communication link, the downstream optical fiber 220 isconnected to a DWDM demultiplexer and the upstream optical fiber 225 isconnected to a DWDM multiplexer. The DWDM MUX/DMUX pair supports Nchannels 240. Each channel 240 has a channel transmitter (TX) 250 and achannel receiver (RX) 260. Each channel transmitter 250 is capable ofcombining K subchannel signals 280 wherein each subchannel is capable ofcarrying a data stream. Conversely, each channel receiver 260 is capableof receiving a channel signal {λ_(i)} and recovering the K data streams290.

Transmitter Design

FIG. 3 a is a block diagram of a transmitter 250 in one embodiment ofthe present invention. The indices used to describer the transmitter inFIG. 3 a correspond to a transmitter for channel i. The transmitter 250comprises a total of K subchannel transmitters 310, each of whichoutputs a data-modulated subchannel signal s_(ik), where ‘k’ refers tothe index of the subchannel transmitter 310 within transmitter 250 forchannel i. A K:1 combiner 320 combines the subchannel signals 342 fromthe K subchannel transmitters into the channel signal 245, {λ_(i)}. Eachsubchannel transmitter 310 includes a subchannel light source 330 and adata modulator 340.

The subchannel light source 330 outputs a subchannel light beam 332having a subchannel wavelength λ_(ik). The subchannel light source maybe either a continuous wave (CW) light source, or a pulsed light beam.Preferably, the wavelength λ_(ik) of the light source 330 is monitoredby a frequency calibrator 600 that controls the light source 330 using afrequency calibration control signal on control line 334. In a preferredembodiment, the K subchannel light sources 330 for channel i arecontrolled by a single frequency calibrator 600 to generate light beamsthat are evenly spaced across the channel bandwidth. For example, in anembodiment where K=16 and the channel bandwidth for channel i is 100GHz, the frequency calibrator controls the K subchannel light sources togenerate the K light beams with a 6.25 GHz frequency spacing betweeneach adjacent subchannel.

The K subchannel light sources 330 for a single channel transmitter 250may comprise K discrete lasers, each of which is individually andindependently controlled by the frequency calibrator 600. Alternatively,K subchannel light sources 330 may be implemented using a single laserwhose output is split and shifted, as needed, to create the K individualsubchannel wavelengths λ_(ik). In this latter implementation, a singlefrequency calibrator controls the output of the single laser and therebycalibrates all subchannel wavelengths λ_(ik) simultaneously in anon-independent manner. Regardless of the nature of the light source330, the subchannel light beam 332 is ultimately directed into a datamodulator 340 where the light beam is modulated by data 370 to producethe subchannel signal 342.

Optionally, the subchannel light beam 332 may be subjected to a pulseshaper 335, which may help optimize the pulse shape for minimum overalldistortion. The pulse shaper 335 itself may be implemented as aMach-Zehnder optical device under control of a transmitter processor(not shown). The resulting pulse shape may belong to a predetermined setof shapes, the particular one chosen at any given time being based onchannel condition information obtained through out-of-band signaling.Alternatively, an arbitrary pulse shape may be adaptively generatedbased on the channel condition information. It is therefore contemplatedthat not only are predetermined pulse shapes, such as Gaussian,symmetric solitons, and the like, possible, but also a host of otherempirically derived ones, as well. In this manner, the spectral shape,duty cycle and other factors may be adjusted to suit channel conditions.

A modulator calibration unit 345 may be used to help ensure that theenergy in the modulated signals λ_(ik) is substantially even among the Ksubchannels. The modulation calibration unit measures the optical energyoutput from each of the data modulators 340, and provides feedback tothe data modulators so as to ensure that all K subchannels havesubstantially similar intensities and also ensure that the members ofthe constellations are orthogonal and balanced. In one sense, then, themodulator calibration unit acts as a power balancer. In a preferredembodiment each of N channels has a single calibration unit 345 thattakes into account only the K subchannels, and this is implemented withthe help of a processor, or the like. It is understood, however, that asingle modulator calibration unit may likewise be employed to serve allN channels, given sufficient processing power. The modulator calibrationunit preferably includes optical detectors which receive input from thechannel signal 342 output by the data modulator, along with circuitry toconvert the detected optical signals into digital form, and a processorto perform any needed calculations and output control signals back tothe data modulator.

FIG. 3 b is a block diagram of a second embodiment of a transmitter1400. The transmitter 1400 employs a single light source 1402 that putsout a plurality of evenly spaced-apart spectral lines, separated by somefrequency spacing Δf. Preferably, the single light source 1402 puts outK such spectral lines, each being used as a subchannel light beam 332.This allows one to employ a single light source 1402 to provide theilluminating beam. The evenly spaced-apart spectral lines in thefrequency domain, may be regarded as the teeth of a “comb”, and so sucha light source may be referred to as a ‘comb generator’. U.S. Pat. No.4,096,448 to Hayes and U.S. Pat. No. 5,347,525 to Fans, whose contentsare incorporated to the extent necessary to understand the presentinvention, exemplify systems for creating optical signals having evenlyspaced-apart frequencies.

In the transmitter 1402 of FIG. 3 b, the comb generator 1402 preferablyincludes circuitry to ensure even frequency spacing between the teeth.Such circuitry therefore effectively serves to calibrate the teethoutput by the comb generator. The output of the comb generator 1402 maythen be subject to a pulse shaper 335 and then modulated with data, ineach of the sub-channel transmitters.

Data Modulator

FIG. 4 is a block diagram of a preferred embodiment of a data modulator340 of the sort used in the subchannel transmitters seen in FIGS. 3 aand 3 b. An incoming subchannel light beam 332 from the subchannel lightsource 330 enters the data modulator 340 through a power splitter 810.The power splitter 810 splits incoming beam 332 into two beams,designated as H and V. The H and V beams are each directed into separate1:2 splitters 820 a, 820 b, although a 1:4 splitter may be used instead.The splitter 820 a splits the H beam into H1 and H2 components anddirects one of these (H2 in FIG. 8), into a 90° phase shifter 825 a.Similarly, the splitter 820 b splits the V beam into V1 and V2components and directs one (V2 in FIG. 8) into a 90° phase shifter 825b.

Each of the four component beams are directed into a separate modulator830 a, 830 b, 830 c, 830 d where the data stream 370 is modulated on thefour component beams. As discussed above, the data stream DATA_(ijk)includes data to be encoded on the H and V polarizations of a singlecode word, depending on the index ‘j’. Thus, the data input tomodulators 830 a, 830 b is the data corresponding to j=‘H’ and the datainput to modulators 830 c, 830 d is the data corresponding to j=‘V’.

Control signals from a modulation calibration unit 345 ensure that theintensity of each data-modulated component beam is comparable to that ofthe others and ensures the orthogonality of the in-phase and quadraturecomponents of the constellation. Thus, the modulation calibration unitmay have inputs into one or both of the phase shifters 825 a, 825 b andthe data modulators 830 a, 830 b, 830 c, 830 d.

In a preferred embodiment, each of the four component beams is modulatedwith one bit. The modulators 830 a, 830 b, 830 c, 830 d are eachimplemented as a Mach-Zehnder Interferometer (MZI), each applying anamplitude of 1 or −1 (0° or 180° phase shift) to the component beamdepending on whether the bit state is a 0 or 1. In a preferredembodiment, the data is encoded in the changes of phase between one bitto the other, referred to as differential quadrature phase shift keying(DQPSK). As will be clear to one of skill in the art, the 90° shift maybe applied after the bit modulator 830. As will also be clear to one ofskill in the signal processing arts, the modulator 830 may also be aphase modulator or a multilevel amplitude modulator thereby allowing anyamplitude/phase modulation and encoding of a higher number of bits persymbol.

The modulated H1 and H2 component beams are combined in a first combiner840 a to form a data-modulated beam H′. Similarly, the modulated V1 andV2 component beams are combined in a second combiner 840 b to form adata modulated beam V′. The data-modulated beams H′ and V′ are thencombined in a polarization beam combiner 850 that rotates thepolarization plane of the H′-beam 90° to the polarization plane of theV′-beam and combines both beams into the data beam 342. The combined H′and V′ beams do not interfere because of their orthogonal polarizationplanes.

As will be clear to one of skill in the optical arts, the polarizationstate of H′ and V′ beams may also be converted to any two orthogonalpolarizations (such as right circularly polarized beam and a leftcircularly polarized beam) before combining into the data beam and stillmaintain the orthogonality condition that prevents interference betweenthe two beams. As will also be clear to one of skill in the opticalsignal processing arts, the bit modulators 830 a, 830 b, 830 c and 830 dmay also be implemented as a phase modulator or a multilevel amplitudemodulator, thereby allowing higher bits per symbol.

Channel Receiver with Fully Digital PMD Compensation

FIG. 5 a is a block diagram of a first embodiment of a channel receiver500, of the sort that can be used as the receiver 260 in FIGS. 2 a & 2b. The nomenclature used to describe receiver 500 corresponds to areceiver for channel i.

The channel signal {λ_(i)} from the DWDM demultiplexer 230, whichcarries K subchannel signals each having a data stream with both H and Vcomponents, is directed into a 1:K receiver splitter 502 that splits theoptical channel signal into K identical receiver channel signals 504.Each of the identical receiver channel signals 504 is input to one of Ksubchannel receivers 510, each subchannel receiver 510 being configuredto detect one subchannel {λ_(ik)}. For purposes of clarity, only onesubchannel receiver 510 is shown in detail in FIG. 5 a.

The receiver channel signal 504 is first directed into a polarizationbeam splitter 540 which splits the incoming signal into two orthogonalcomponents, designated ‘H’ and ‘V’. The H- and V-components aresubjected to variable delays 550 a, 550 b to form delayed componentH_(d), V_(d), respectively. Both components are subjected to delayssince it may not be known in advance which of the two, if any, is aheadof the other and the so the delay assists in timing synchronizationbetween the two. The delayed H-component H_(d) is directed as a firstinput into a first optical phase detector 700 a while the V-componentV_(d) is directed as a first input into a second optical phase detector700 b.

A local subchannel light source 530 is configured to output a referencesignal having a subchannel wavelength, λ_(ik). The subchannel lightsource 530 preferably is implemented in the same manner as thesubchannel light source 330 associated with the transmitter seen in FIG.3 a. More preferably, the same unit 330 is used since the transmitterand receiver are co-located. A frequency calibrator 600 is thus alsoused to output a frequency calibration control signal 534 to control thelocal subchannel light source 530.

The output from the local subchannel light source 530 is input to a 1:2splitter 545 to thereby output two identical reference signals 547 a,547 b. Of these, reference signal 547 a is directed as a second input tothe first optical phase detector 700 a, while reference signal 547 b isdirected as a second input to the second optical phase detector 700 b.And since channel receiver 500 mixes the incoming signal with a localreference signal which preferably is at the same frequency, channelreceiver 500 can be considered to be a homodyne receiver.

The optical detector 700 a combines the delayed H-component H_(d) withreference signal 547 a and outputs a digitized electrical signal 452 a,indicated as h′[p] in FIG. 5 a. Similarly, optical detector 700 bcombines the V-component with reference signal 547 b and outputs adigitized electrical signal 452 b, indicated as v′[p] in FIG. 4 a.Digital electrical signals 452 a and 452 b each include the separate Iand Q components. The ‘prime’ notation for h′[p] and v′[p] indicatesthat the digitized signals 452 a, 452 b are uncompensated for anypolarization distortion and the ‘p’ refers to an index in a continuum ofdigitized signals.

The digitized electrical signals 452 a, 452 b are then input to adigital processor 560 associated with the receiver for furtherprocessing. The processor may be realized by a general purposemicroprocessor, an FPGA, an ASIC, a specialized RISC architecture, orthe like. Regardless of how the processor is implemented, it isconfigured to perform a number of tasks including data demodulation,symbol timing recovery, and PMD compensation, which in the embodiment ofFIG. 5 a is completely by digital manipulation of the signals. Each ofthese is implemented as some combination of hardware, firmware and/orsoftware modules associated with the processor.

FIG. 5 c shows a functional diagram of the some of the criticalprocessing performed by the processor 560.

With regard to PMD compensation, the processor 560 is provided with aPMD compensation module 591 which monitors incoming first and seconddigital signals 452 a, 452 b (h′[p] & v′[p]) and internally produces thePMD-compensated first and second digital signals h[p] & v[p] therefrom.In principle, the processor 560 digitally aligns the SOP of the receivedsignal to the SOP of the transmitted signal. This is performed by arotation operation given by the matrix equationR[p]=TR′[p]  (1)whereR′[p]=(v′[p]h′[p])^(T)  (2)R[p]=(v[p]h[p])^(T)  (3)T is a 2×2 rotation matrix, the superscript T represents the transposeand h[p] and v[p] (without the ‘prime’) represent the compensateddigitized electrical signals. Preferably, this calculation is donedigitally within the processor 560.

With regard to data demodulation, the processor 560 is provided with adata demodulation module 592. The data demodulation module uses theinternal polarization-compensated digital signals h[p] and v[p] torecover the data which modulated the subchannel wavelength, λ_(iK). Thereceiver then outputs the subchannel data stream Data_(ijK).

The data demodulation module 592 also incorporates a frequency offsetcompensator. As discussed above, the subchannel receiver 520 uses alocal reference laser 530 to convert the optical signal into theelectrical domain. The frequency difference between the incoming carriersignal and the local reference laser 530 produces cross talk between thein-phase and quadrate components in the subchannel receiver and degradethe phase detection. To minimize degradation due to this frequencyerror, the laser source at the transmitter and receiver have to betightly calibrated. The calibration requirements can be relaxed if theresidual carrier frequency error is compensated for down in the receiverchain. This may potentially allow one to use a simpler system withrespect to ensuring identical transmitter and receiver frequencies.

Thus, two approaches to transmitter/receiver frequency correspondencemay be achieved—the first approach using a frequency calibration unit600 described below, and the second approach using a frequency offsetcompensator associated with data demodulator module 592. This secondapproach helps mitigate additional frequency offsets that may affect thetransmitted signal as it propagates through the channel. Under certainconditions, it is contemplated that one may be able to altogether doaway with the frequency calibration circuit, and use only the frequencyoffset compensator.

The frequency offset compensator within the data demodulator module 592estimates the frequency error and performs the compensation either byrotating the received waveform, or, in case of the differentiallyencoded signal, by shifting the signal phase (or biasing the decisionboundary). Depending on the modulation format, maximum allowablefrequency offset, noise level, etc., an appropriate frequency offsetestimation algorithm can be chosen from among the standard algorithmsknown to those skilled in the art. U.S. Pat. No. 5,245,611, whosecontents are incorporated by reference, discloses such an algorithmwhich may be readily adapted from the wireless domain to the opticaldomain. In a preferred embodiment, a decision-directed frequencyestimation and phase shift compensation algorithm is employed.

With regard to symbol timing recovery, the processor 560 is providedwith a symbol timing recovery module 593 that monitors the digitalsignals 452 a, 452 b, and also their internal PMD-compensated versionsh[p] and v[p], and outputs a suite of optical detector control signals,represented by 581 to the optical detectors 700 a, 700 b. The suite ofcontrol signals 581 is used for symbol timing recovery, synchronizationof analog and digital components, and the like. Included among thesecontrol signals are timing signals to control the sampling of theoptical detectors. The optical detectors 700 a, 700 b require thesesignals to indicate the data symbol temporal boundaries in the opticalchannel signal. If the timing signal is not aligned to the symbolboundary, the detected signal will include additional noise from theother subchannels thereby increasing the bit error rate of the signal(it would also include inter-symbol interference noise). The normalizedtiming error, τ, is the difference between the timing signal that beginsthe sampling in the optical detector 700 a, 700 b and the actual symbolboundary in the optical channel signal divided by the symbol period.

The processor 560 also outputs a clock signal 583 to control thevariable delays 550 a, 550 b. While in FIG. 5 a, the clock signal isshown as coming directly from the processor 560, it may instead becreated by a separate device, perhaps controlled by the processor 560.

A receiver controller module 594 supervises the operation of some of theother modules in processor 560.

Channel Receiver with Driver for Mechanical Polarization Compensation

FIG. 5 b is a block diagram of a second embodiment of a channel receiver570 of the sort that can be used for receiver 260 of FIGS. 2 a and 2 b.The channel receiver 570 includes K subchannel receivers 520 which haveslightly different construction than subchannel receivers 510 seen inFIG. 5 a. However, channel receiver 570 also mixes the incoming signalwith a local reference signal which preferably is at the same frequency,and so channel receiver 570 can also be considered to be a homodynereceiver.

The channel signal, {λ_(i)}, from the DWDM demultiplexer 230 is directedto the 1:K receiver splitter 502 that splits the optical channel signalinto K identical receiver channel signals 504, each of which is directedinto a subchannel receiver 520. For purposes of clarity, only onesubchannel receiver 520 is shown in detail in FIG. 5 b.

The channel signal, {λ_(i)}, entering the subchannel receiver 520 isfirst directed to an optical PMD compensator device 538 (RX PC) and ismodified into a polarization-compensated beam 506. The PMD compensator538 adjusts the SOP of the channel signal {λ_(i)} to align the receivedsignal's SOP to the SOP of the transmitted signal and thereby compensatefor PMD distortion caused by the transmitting medium such as the opticalfiber. The PMD compensator 538, in one embodiment, is the Acrobat™Polarization Control Module (PCM) from Corning Incorporated of Corning,N.Y. The PolarRITE™II Polarization Controller from General PhotonicsCorporation of Chino, Calif. may also be used as the PMD compensator538.

The polarization-compensated beam 506 is first directed into apolarization beam splitter 540 that splits the incoming signal into twoorthogonal components, designated ‘H’ and ‘V’. The H-component isdirected as a first input into a first optical phase detector 700 awhile the V-component is directed as a first input into a second opticalphase detector 700 b.

A local subchannel light source 530 is configured to output a referencesignal having a subchannel wavelength, λ_(ik). The subchannel lightsource 530 preferably is implemented in the same manner as thesubchannel light source 330 associated with the transmitter seen in FIG.3 a and, more preferably, is the same unit since the transmitter andreceiver are co-located. A frequency calibrator 600 is thus also used tooutput a frequency calibration control signal 534 to control the localsubchannel light source 530.

The output from the local subchannel light source 530 is input to a 1:2splitter 545 to thereby output two identical reference signals 547 a,547 b. Of these, reference signal 547 a is directed as a second input tothe first optical phase detector 700 a, while reference signal 547 b isdirected as a second input to the second optical phase detector 700 b.

The optical detector 700 a combines the H-component with referencesignal 547 a and outputs a digitized electrical signal 454 a, indicatedas h[p] in FIG. 5 b. Similarly, optical detector 700 b combines theV-component with reference signal 547 b and outputs a digitizedelectrical signal 454 b, indicated as v[p] in FIG. 5 b. Digitalelectrical signals 454 a and 454 b each include the separate I and Qcomponents. By virtue of the effect of the PMD compensator 538, thesignals h[p] and v[p] (which do not include the ‘prime’ notation) arethe polarization-compensated data modulated digital signals, with ‘p’referring to an index in a continuum of digitized signals.

The digitized electrical signals 454 a, 454 b are then input to adigital processor 575 associated with the receiver 570 for furtherprocessing. The processor 575 may be realized by a general purposemicroprocessor, an FPGA, an ASIC, a specialized RISC architecture, orthe like. Regardless of how the processor 575 is implemented, it isconfigured to perform a number of tasks including data demodulation,symbol timing recovery, and PMD compensation control with the opticalcompensation being performed by a physical device. Each of these isimplemented as some combination of hardware, firmware and/or softwaremodules associated with the processor.

FIG. 5 d shows a functional diagram of the some of the criticalprocessing performed by the receiver processor 575.

With regard to data demodulation, the processor 575 is provided with adata demodulation module 595 that monitors the PMD-compensated digitizedsignals 454 a, 454 b and recovers the data that modulated the subchannelwavelength λ_(ik) The receiver 575 then outputs the subchannel datastream Data_(ijK). The data demodulation module 595 in receiverprocessor 575 also incorporates a frequency offset compensator, muchalong the lines described above with respect to the one in receiverprocessor 560.

With regard to PMD compensation, the receiver is provided with a PMDcompensation control module 596 that monitors the digitized signals 454a, 454 b and generates PMD device control signals 585 that are providedto the PMD compensator 538 based on these signals. The PMD compensationcontrol module continually adjusts the channel signal's SOP through thePMD compensator 538 to find and track the optimum adjustment for the PMDcompensator 538. The criteria for optimum adjustment of the PMDcompensator 538 may be based on maximizing the SNR of the digitizedsignals 454 a, 454 b or other such criteria known to those skilled inthe art of optical signal processing.

With regard to symbol timing recovery, the receiver processor 575 isprovided with a symbol timing recovery module 597 that monitors thepolarization-compensated digitized signals 454 a, 454 b and outputs asuite of optical detector control signals 581 to the optical detectors700 a, 700 b. The suite of control signals 581 is used for symbol timingrecovery, synchronization of analog and digital components, and thelike. Included among these control signals are timing signals to controlthe sampling of the optical detectors. The optical detectors 700 a, 700b require these signals to indicate the data symbol temporal boundariesin the optical channel signal. If the timing signal is not aligned tothe symbol boundary, the detected signal will include additional noisefrom the other subchannels thereby increasing the bit error rate of thesignal (it would also include inter-symbol interference noise). Thenormalized timing error, τ, is the difference between the timing signalthat begins the sampling in the optical detector 700 a, 700 b and theactual symbol boundary in the optical channel signal divided by thesymbol period. For the receiver processor 575, the symbol timingrecovery module preferably is implemented as a Mueller & Muller timingerror detector module, discussed below.

The processor 575 also outputs a clock signal 583 to control thevariable delays 550 a, 550 b. While in FIG. 5 b, the clock signal isshown as coming directly from the processor 575, it may instead becreated by a separate device, perhaps controlled by the processor 575

Finally, a receiver controller module 598 supervises the operation ofthe other modules in processor 575.

Self-Homodyne Sub-Channel Receiver

The sub-channel receivers 510, 520 described above utilize a localsubchannel light source 530, and so may be regarded as ‘heterodyne’ ifthe transmit and receive frequencies of the lasers are different and‘homodyne’ if they are the same of close to one another. As discussedabove, one difficulty with such receivers is the problem of frequencyoffset between the transmitter light source and the receiver lightsource. One way to obviate the problem of frequency offset is to use aself-homodyne subchannel receiver which does not need a local subchannellight source.

FIGS. 11 a, 11 b and 11 c show different embodiments of self-homodynereceivers 1100 a, 1100 b, 1100 c in accordance with the presentinvention. The self-homodyne receivers 1100 a, 1100 b, 1100 c eachinclude a 1:K optical demultiplexer 1102 which receives themulti-frequency channel signal {λ_(l)} and outputs the K individualsub-channel signals {_(lk)}, shown on signal path 1104. The opticaldemultiplexer 1102 thus performs the equivalent of optically splittingthe multi-frequency channel signal into K separate paths and thenemploying an optical filter attuned to a particular frequency in each ofthose paths. Each of the self-homodyne receivers 1100 a, 1100 b, 1100 ccomprises K self-homodyne sub-channel receivers 1110 a, 1110 b, 1110 c,respectively. Since all of the self-homodyne sub-channel receivers forany given self-homodyne receiver 1100 a, 1100 b, 1100 c are the same,only one of each is described herein.

FIG. 11 a shows a first embodiment of a self-homodyne sub-channelreceiver system. This self-homodyne embodiment has a single PMDcompensator and a single polarization beam splitter. As seen in FIG. 1a, the sub-channel signal 1104 is input to a PMD compensator device 1138in self-homodyne sub-channel receiver 1110 a, resulting in aPMD-compensated signal 1106. PMD compensator device 1138 is controlledby control signals 1185 from the receiver processor 1175, not unlike thesub-channel receiver 520 described above. The PMD-compensated signal1106 is then input to a polarization beam splitter 1140 which outputsthe H- and V-polarized signals 1108 h, 1108 v, respectively.

The H-polarized signal 1108 h is input to 1:2 splitter 1142 a to producetwo identical H-polarized signals 1108 h, one of which is delayed by aone-symbol delay 1144 a to produce a delayed H-polarized signal 1109 h.The H-polarized signal 1108 h and the delayed H-polarized signal 1109 hare then input to a polarization-sensitive optical detector 1150 aoutputting h_(d)[p] as signal 1170 h. h_(d)[P] comprises a digitaloutput signal representative of both the I and Q components of phaseand/or amplitude difference between corresponding bits of information insuccessive symbols encoded on H polarization at the transmitter.

The V-polarized signal 1108 v is subject to a similar treatment, beingpassed through 1:2 splitter 1142 b, one output of which is subjected todelay 1144 b to produced delayed V-polarized signal 1109 v. DelayedV-polarized signal 1109 v and V-polarized signal 1108 v are input to apolarization-sensitive optical detector 1150 b which outputs V_(d)[p] assignal 1170 v. V_(d)[p] comprises a digital output signal representativeof both the I and Q components of phase and/or amplitude differencebetween corresponding bits of information in successive symbols encodedon V polarization at the transmitter.

As is known to those skilled in the art, a polarization sensitive deviceis one having optical circuitry that preserves the polarization of anoptical signal input thereto. This is usually achieved through selectionof specific materials, such as lithium niobate, LiNbO₃, which areparticularly well-suited for this purpose. Thus, while the opticalcircuitry of a polarization-sensitive and polarization-insensitivedevice may be similar, the material used to form the crystal supportingthe waveguides and other structures may differ.

The receiver processor 1175 provides much of the same functionality asreceiver processor 575. Thus, receiver processor 1175 includes analogousmodules and also outputs optical detector control signals 1181, symboldelay signals 1183 and PMD device control signals 1185. One significantdifference, however, is in the nature of the data demodulation moduleassociated with receiver processor 1175. Since the signals 1170 h, 1170v are differential data, such as from DPSK modulation, the datademodulation module associated with receiver processor 1175 treats thesedata differently than does data demodulation module 595 associated withreceiver processor 575.

FIG. 11 b presents the second embodiment of a self-homodyne receiver1100 b having self-homodyne sub-channel receivers 1110 b. Thisself-homodyne embodiment has a single PMD compensator and multiplepolarization beam splitters. In self-homodyne sub-channel receiver 1110b, The sub-channel signal 1104 is input to a PMD compensator device1138, resulting in a PMD-compensated signal 1106. Again, the PMDcompensator device 1138 is controlled by control signals 1185 from thereceiver processor 1175. The PMD-compensated signal 1106 is then inputto 1:2 splitter 1142 to produce two identical PMD-compensated signals1152 a, one of which is delayed by a one-symbol delay 1144 to produce adelayed PMD-compensated signal 1152 b.

The PMD-compensated signal 1152 a and the delayed PMD-compensated signal1152 b are then input to a polarization-sensitive 90° optical hybridcircuit 1200 which outputs four combined signals 731, 735, 733 and 737,discussed further below with respect to the optical 900 hybrid detector799 seen in FIG. 7. The polarization-sensitive 90° optical hybridcircuit 1200 is similar to the optical 90° hybrid detector 799, but doesnot include the two pairs of matched photodiodes. The four combinedsignals 731, 733, 735 and 737 represent the four complexcross-components of the PMD-compensated signal 1152 a and the delayedPMD-compensated signal 1152 b.

The four combined signals 731, 735, 733 and 737 are input to respectivepolarization beam splitters 1140 a, 1140 b, 1140 c and 1140 d,respectively, to produce four corresponding pairs of H- and V-polarizedsignals designated 731 h, 735 h, 733 h and 737 h, and 731 v, 735 v, 733v and 737 v in FIG. 11 c.

H-polarized signals 731 h and 735 h are input to a first matcheddetector 1162 a which includes light sensors, such as photodiodes, tothereby generate analog electrical detection signals 1164 a proportionalto the intensity difference between the H-polarized signals 731 h and735 h. Similarly, H-polarized signals 733 h and 737 h are input to asecond matched detector 1162 b to produce analog electrical detectionsignals 1164 b, V-polarized signals 731 v and 735 v are input to a thirdmatched detector 1162 c to produce analog electrical detection signals1164 c and V-polarized signals 733 v and 737 v are input to a fourthmatched detector 1162 d to produce analog electrical detection signals1164 d.

The analog electrical detection signals 1164 a, 1164 b corresponding tothe H-polarization are input to a first post-detection circuit 1300 aseen in FIG. 13 as post-detection circuit 1300. First post-detectioncircuit 1300 a outputs h_(d)[p] as signal 1170 h which comprises adigital output signal representative of both the I and Q components ofphase and/or amplitude difference between corresponding bits ofinformation in successive symbols encoded on H polarization at thetransmitter. Similarly, the analog electrical detection signals 1164 c,1164 d corresponding to the V-polarization are input to a secondpost-detection circuit 1300 b which outputs v_(d)[p] as signal 1170 v,and is the V-polarization counterpart.

The receiver processor 1175 in FIG. 11 b receives and processes hd[p]and V_(d)[p] as discussed above, outputting the various control signals1181, 1183, 1185 and also the data processed by sub-channel receivers1110 b.

FIG. 11 c presents the third embodiment of a self-homodyne receiver 1100c having self-homodyne sub-channel receivers 110 c. This self-homodyneembodiment has a multiple PMD compensators and multiple polarizationbeam splitters. In self-homodyne sub-channel receiver 1110 c, Thesub-channel signal 1104 is first input to a 1:2 splitter 1142 to producetwo identical sub-channel signals 1104 a, one of which is delayed by aone-symbol delay 1144 to produce delayed sub-channel signal 1105 a.

The sub-channel signal 1104 a and the delayed sub-channel signal 1105 aare input to a 90° optical hybrid circuit 1200 which outputs the fourcombined signals 731, 735, 733 and 737. The 90° optical hybrid circuit1200 found in sub-channel receiver 1110 c does not have to bepolarization sensitive, since it does not receive the polarizationbeam-split signals. However, it would not hurt to use apolarization-sensitive 90° optical hybrid circuit. The four combinedsignals 731, 733, 735 and 737 represent the four complexcross-components of the PMD-compensated signal 1152 a and the delayedPMD-compensated signal 1152 b.

The four combined signals 731, 735, 733 and 737 are each input to acorresponding PMD compensator device 1138 a, 138 b, 1138 c and 1138 d,respectively, to form PMD-compensated combined signals 731 c, 735 c, 733c and 737 c, respectively. Each of the PMD compensator devices 1138 a,138 b, 1138 c and 1138 d is controlled by the suite of signals 1185 fromthe receiver processor 1175, created by the PMD compensation controllermodule therein.

The four PMD-compensated signals 731 c, 735 c, 733 c and 737 c are inputto a corresponding polarization beam splitter 1140 a, 1140 b, 1140 c and1140 d, each outputting a pair of signals having first and secondorthogonal polarizations. The remainder of the circuitry of thesub-channel receiver 1110 c is substantially the same as that ofsub-channel receiver 1110 b described with respect to FIG. 11 b. Thereceiver processor 1175 in the circuit of FIG. 11 c receives andprocesses h_(d)[p] and v_(d)[p] as discussed above, outputting thevarious control signals 1181,1183, 1185 and also the data processed bysub-channel receivers 1110 c.

An optical communications system may employ either of the transmitters250, 1400 in conjunction with either receivers 500, 570 or theself-homodyne receivers 1100 a, 110 b, 1100 c. This is because theself-homodyne technique of comparing a signal with a one-symbol delayedversion of itself does not depend on the nature of the transmittedsignal.

Frequency Calibrator

FIG. 6 is a block diagram of the frequency calibrator 600 in a preferredembodiment of the present invention. The K light beams, eachrepresenting a subchannel frequency λ_(ik) from either the transmittersubchannel light source 330 or from the receiver subchannel light source430 are directed into a K×1 switch 610. The K×1 switch 610 selects oneof the subchannel frequencies λ_(ik) for calibration and directs thechosen light beam 611 to a second switch 620, which is a 2×1 switch. Itshould be noted, however, that a K+1:1 switch may be used instead offirst and second switches 610, 620. Second switch 620 selects eitherchosen light beam 611 or reference light beam 622 generated by referencelaser 621 and outputs a selected beam 625. Thus, switches 610, 620collectively, or the single K+1:1 switch by itself, serve as an opticalan optical switch system configured to select one from among the K laserlight sources and a reference beam and output the selected beam 625.

The selected beam 625 is directed into a 1:2 splitter 630 to createidentical first and second selected split beams. One of the selectedsplit beams is directed as a first input 642 into an optical detector650. The second selected split beam is directed into a delay 640 beforebeing directed as a second input 644 into the optical detector 650.Preferably, the delay 640 comprises a delay line having a lengthadjusted such that the delay applied to the second split beam is onesymbol period. The optical detector 650 converts the two optical signalsinto a digitized electrical signal 652 that is proportional to e^(jωT),where T is the delay of delay line 640 and ω is the frequency of theselected beam 625.

A controller 600 is configured to receive the digitized electricalsignal 652 and output one more frequency calibration control signals tocontrol the light source(s) responsible for creating the K laser lightbeams 601. More particularly, controller 660, which preferably isimplemented as a microprocessor or the like, receives the digitizedsignal 652 from the optical detector 650 and estimates the frequency ofthe selected beam 625 using the known value of the delay, T. Thecontroller 660 compares the measured frequency to the desired frequency,ω_(ik), and generates frequency calibration control signal 334, 534,which is sent to the subchannel light source corresponding to the chosenlight beam 611. The frequency (wavelength) of the subchannel lightsource is thus adjusted to the desired frequency based on the frequencycalibration control signals 334, 534.

The empirical value of the delay, T, is calculated by selecting thereference light beam 622 and measuring the phase of the resulting twosplit beams. The reference laser 621 is preferably a CW laser that iscapable of generating a very stable light beam of known frequency. Thecontroller 660 determines T from signal 652 based on the known value ofthe reference laser frequency. The value of T determined by thecontroller 660 is stored and used by the controller 660 to measure thefrequencies of the subchannel light beams 601. The controller outputs asuite of optical detector control signals 681 and also controls switches610, 620 such that each subchannel light source is sequentiallycalibrated.

In a preferred embodiment a single frequency calibrator is used for thek subchannels and is used for both the transmitter and the receiver,with each pair of the co-located transmitters 250 and receivers 260sharing that single frequency calibrator. Alternatively, a singlefrequency calibrator may be used for all N channels, with all Nco-located transmitters 250 and receivers 260 sharing that singlefrequency calibrator. In such case, the single frequency calibrator istime-shared by the N channels through a switching mechanism whichselects the light source associated with a particulartransmitter/receiver pair being calibrated at any given instant.

Optical Phase Detector

FIG. 7 presents a block diagram of the optical phase detectors 700, seenin FIGS. 5 a and 5 b and the optical phase detector 650 seen in FIG. 6.The optical phase detector receives two optical beams 705, 701 andgenerates two electrical signals 795, 791 corresponding to the in-phaseand quadrature components of the multiplication of 705 by the complexconjugate of 701. One of the two beams, designated as A in FIG. 7, isreferred to as a signal beam. The second optical beam 701, designated asB in FIG. 7, is referred to for convenience as a reference beam in thisdescription of the optical phase detector. It should be kept in mind,however, that the ‘reference beam’ 701 may be something other than anunmodulated train of pulses having no information.

The signal beam 705 is split into four beams identified by 715 in FIG.7. In a referred embodiment, the signal beam 705 is split into fourbeams by a cascade of 1:2 splitters 710. In another embodiment, thesignal beam 705 may be directly split into four beams by a single 1:4splitter. Regardless of how they are formed, the four signal beams 715are directed into four 2:1 combiners 730 a, 730 b, 730 c, and 730 d.

The reference beam 701 is first subjected to a 1:2 splitter 720 a toform identical beams R1 and R2. Beam R1 is then subjected to a second1:2 splitter 720 b to form identical beams R3 and R4. Meanwhile, Beam R2is first subjected to a 180° phase shifter 724 before being spilt by 1:2splitter 720 c to thereby form identical beams R5 and R6. Beam R3 isinput as signal 721 to combiner 730 a while beam R4 is first subjectedto a first 90° phase shifter 722 a before being input to combiner 730 bas signal 723. Beam R5 is input as signal 725 to combiner 730 c whilebeam R4 is first subjected to a second 90° phase shifter 722 b beforebeing input to combiner 730 d as signal 727.

The resulting beams R3, R4, R5 and R6, represented as signals 721, 723,725, and 727 are phase-shifted by 90° increments. Beam R3/721 has zerophase shift and is combined with one of the four signal beams 715 in 2:1combiner 730 a to produce first combined beam 731 that is the sum of thesignal and reference beam, designated as A+B. Beam R4/723 has a 90°phase shift and is combined with one of the four signal beams 715 insecond 2:1 combiner 730 b to produce second combined beam 733 that isdesignated as A+jB. Beam R5/725 has a 180° phase shift and is combinedwith one of the four signal beams 715 in third 2:1 combiner 730 c toproduce third combined beam 735 that is the difference between thesignal beam and reference beam and is designated as A−B. Finally, beamR6/727 has a 270° phase shift and is combined with one of the foursignal beams 715 in fourth combiner 730 d to produce fourth combinedbeam 737 that is designated as A−jB.

The first and third combined beams 731, 735 are input to a first matcheddetector 740 a. The first matched detector 740 a includes light sensors745 a, to thereby generate electrical signals 798 a that areproportional to the intensity difference between the first and thirdcombined beams 731, 735. The light sensors 745 a are preferablyphotoelectric detectors such as p-n, p-i-n, or Schottky-barrierphotodiodes, and are selected to generate substantially identicalelectrical signals for the same incident light beam. The electricalsignals 798 a generated by the first matched detector 740 a are input toa first amplifier 750 a. The output signal 752 of the first amplifier750 a is proportional to the in-phase difference between the signal beam705 and the reference beam 170 (real{AB*}) and so can be considered adetected analog in-phase signal.

The second and fourth combined beams 733, 737 are directed to a secondmatched detector 740 b comprising matched light sensors 745 b, tothereby generate electrical signals that are proportional to theintensity difference between the second and fourth beams 733, 737. Theelectrical signals 798 b generated by the second matched detector 740 bare input into a second amplifier 750 b. The output signal 754 of thesecond amplifier 750 is proportional to the quadrature phase differencebetween the signal beam 705 and the reference beam 701 (imag{AB*}) andso can be considered a detected analog quadrature signal. The splitters,phase shifters, combiners and detectors together comprise an optical 90°hybrid detector 799 that outputs the detected analog signals 798 a, 798b. Additional information about the operation and construction of theoptical hybrid detector 799 is described in Kazovsky, et al., “All-fiber90° optical hybrid for coherent communications, ” Applied Optics [23]3,February, 1987.

The analog amplifiers 750 a, 750 b preferably are controlled by gaincontrol signals 772 a, 772 b, respectively, provided by a phase detectorcontroller (not shown in FIG. 7). The amplified detected analog signalsfrom the amplifiers 750 a, 750 b are input to filters 760 a, 760 b,respectively, which are controlled by clock adjustment signals 774 a,774 b, respectively, from the phase detector controller, whichpreferably is implemented as part of a synchronization and symbol timingmodule associated with the receiver's processor.

The filters 760 a, 760 b allow only the subchannel signal correspondingto the λ_(ik) of the local light source to pass through to the sample &holds (S&H) 780 a, 780 b, respectively. In a preferred embodiment forthe detector 650 seen in the frequency calibration unit 600 of FIG. 6,the filters 760 a, 760 b are implemented as analog low pass filters.

In a preferred embodiment for transmitters in which the frequencyspacing between the spectral lines is some value Δf, the filters 760 a,760 b are implemented as integrate & dump (I & D) filters whichintegrate the received signal over one symbol period of duration 1/Δf.The I & D filter is based on a property described by the equation:

$\begin{matrix}{{\Delta\;{f \cdot {\int_{t = 0}^{{1/\Delta}\; f}{{{\mathbb{e}}^{{j2}\;{\pi{({f_{o} + {{k \cdot \Delta}\; f}})}}t} \cdot {\mathbb{e}}^{{- {j2}}\;\pi\;{({f_{o} + {{l \cdot \Delta}\; f}})}t}}\ {\mathbb{d}t}}}}} = \delta_{k,l}} & (4)\end{matrix}$where Δf is the frequency spacing between subchannels, k and I aresubchannel indices, and f_(o) is the channel frequency. The orthogonalproperty described in equation 1 indicates that a reference signal(represented as one of the exponentials) is able to select the onesignal matching the frequency of the reference signal from a multiplexedsignal. In order to make use of the orthogonal property of equation 4,the subchannel frequencies must have a frequency spacing of Δf. In orderto sample only one symbol per integration, the integration period, 1/Δf,must be less than the symbol period. In a preferred embodiment, thefrequency spacing is 6.25 GHz resulting in an integration period of 160ps which is less than the 200 ps symbol period (5.0 GHz symbol rate).The start of integration and the integration period is determined by atiming signal 774 a, 774 b provided to the filters 760 a, 760 b from thesymbol time recovery module within the receiver's processor.

The detected analog in-phase and quadrature subchannel signals output bythe 760 a, 760 b, respectively, is then subject to dc bias adjustment770 a, 770 b based on bias signals 776 a, 776 b, respectively. The dcbias adjustment is provided by the phase detector controller based onthe energy in the signals output by the filters.

The filtered and bias-adjusted analog subchannel signals I′ and Q′,respectively, are converted to digital form by sample and hold (“S&H”)units 780 a, 780 b, respectively, which are controlled bysynchronization signals 778 a, 778 b, respectively from the phasedetector controller.

The output of the sample and hold units is then sent on toanalog-to-digital converters (ADCs) 790 a, 790 b, to form the in-phase I795 and quadrature Q 791 digitized subchannel signals which may becarried on common bus 796. The ADCs 790 a, 790 b are themselvescontrolled by ADC control signals 782 a, 782 b, respectively from thephase detector controller.

PMD Compensation

As discussed above with reference to FIG. 4, each symbol contains fourbits of information from a data stream and is multiplexed onto twoorthogonal polarization states before being transmitted through anoptical fiber to the receiver. The optical fiber causes the propagatingsignal to undergo PMD resulting in a rotation of the originalpolarization states. The rotation is time varying and random anddepends, in part, on the temperature and stress along the fiber. As aresult of the rotation, the SOP of the received signal will not bealigned with the polarization axes of the subchannel polarizing beamsplitter 540 and the beams exiting the polarizing beam splitter willcontain signals from both of the original polarization states instead ofjust one of the original polarization states thereby decreasing the SNRof the received signal. The purpose of the PMD compensation moduleassociated with the receivers 560, 575 is to restore the SOP of thereceived signal to minimize the effect of the PMD caused by the fiber.

FIG. 8 illustrates the problem of PMD and the effect of PMDcompensation. FIG. 8 shows a Poincare sphere representation 870 of thepolarization state of a signal. The SOP of a signal may be representedby a 3-element Stokes vector S=(s₁, s₂, s₃), where s₁ is the differencein the normalized power of the linear horizontal and verticalpolarization components of the signal, s₂ is the difference in thenormalized power of the linear +45° and −45° polarization components ofthe signal, and s₃ is the difference in the normalized power of thecircularly right and left polarization components of the signal. Thepoints R and L on the sphere represent a right circularly polarizedsignal and a left circularly polarized signal, respectively. The pointsH, V, P and Q on the sphere represent linearly polarized horizontal,vertical, 45° and −45° signals, respectively.

The components of the Stokes vector, S, are given by the equations:S=(s ₁ , s ₂ , s ₃)  (5)wheres ₁=(vv*−hh*)/s _(o)  (6)s ₂=2Re(vh*)/s _(o)  (7)s ₃=2lm(vh*)/s _(o)  (8)s _(o)=(vv*+hh*)/s _(o).  (9)The term, s_(o), is a normalizing factor such that all Stokes vectorshave a unit length and the locus of all possible Stokes vectors traces asphere 870 of unit radius. The Stokes components are calculated from anytwo orthogonal polarization states of the signal, designated as v and hin the equations. The SOP of a signal may be transformed by performingrotations about two axes. For example, in FIG. 6 vector s_(a) 810 may betransformed into vector s_(b) 812 by a rotation, θ, about the R-L axis603. Similarly, vector s_(c) 820 may be displaced into vector s_(d) 822by a rotation, ε, about the P-Q axis 802. Thus, the transformation ofany unit vector into another may be described by a pair of rotations, θand ε.

The PMD compensation control module 592, 596 associated with eitherprocessor 560 of FIG. 5 a or processor 575, respectively, of FIG. 5 bcalculates these rotations θ and ε for each polarization component.

In the embodiment corresponding to FIGS. 5 a & 5 c, this is donedigitally by the PMD compensation control module 592 by calculating theoperation given by equations 1–3. For this, it must first calculate theneeded rotations and then form the rotation matrix T. The T matrix hascoefficients T_(rc), where r and c are indices that run from 1 to 2,given by:T ₁₁=cos(θ_(v)/2)cos(ε_(v)/2)−j sin(θ_(v)/2)sin(ε_(v)/2)  (10)T ₁₂=−sin(θ_(v)/2)cos(ε_(v)/2)+j cos(θ_(v)/2)sin(ε_(v)/2)  (11)T ₂₁=sin(θ_(h)/2)cos(ε_(h)/2)+j cos(θ_(h) /2)sin(ε_(h)/2)  (12)T ₂₂=cos(θ_(h)/2)cos(ε_(h)/2)+j sin(θ_(h)/2) sin(ε_(h)/2),  (13)with the angles θ_(v), ε_(v), θ_(h) and ε_(h) representing the rotationof the v′ and h′ polarization components.

In the embodiment corresponding to FIGS. 5 b & 5 d, the PMD compensationcontrol module 596 associated with the processor 575 calculates thesesame angles and outputs them as control signals to drive the PMDcompensator 538.

All-Digital PMD Compensation Algorithm

FIG. 9 a is a flow diagram 900 of the PMD compensation control algorithmwithin the PMD compensation control module 591 associated with receiverprocessor 560. The PMD compensation control algorithm preferably isimplemented in software. The PMD compensation algorithm executes asearch procedure to find the coefficients of the rotation matrix Twhich, when applied to the digitized signals 452 a, 452 b of FIG. 5 a,restores the original SOP of its components. The search of thecoefficients of T is directed towards the optimization of at least onepredetermined criteria. Only the optimal rotation matrix T, asdetermined by the PMD compensation control algorithm, is applied to thedigitized signals 452 a, 452 b comprising h′[p] & v′[p] to create thecompensated signals h[p] & v[p]. Thus, the compensated digital signalsh[p], v[p] do not inherit the noise associated with the search routine.

In a preferred embodiment, the sequence 900 represented in FIG. 9 isperiodically entered at step 910 while the channel receiver 500 receivesoptical signals. The search is an iterative procedure. For eachiteration loop k, the current state of the rotation matrix T[k] isdefined by the two pairs of angles (θ_(v)[k−1], ε_(v)[k−1]) and(θ_(h)[k−1], ε_(h)[k−1]) as in equations 10–13.

In step 915, a candidate step size Z(k), preferably dependent on theloop iteration number k, is determined.

In step 920, the next search step, which includes a step direction, iscalculated. The search step is defined as a pair of incremental angles(Δθ, Δε) that corresponds to the SOP rotation by Δθ and Δε around theR-L and P-Q axes on the Poincare sphere. For each iteration loop k, thesearch procedure investigates nine possible SOP rotation directionsdefined by the angles (θ_(v)[k−1]+mΔθ, ε_(v)[k−1]+lΔε, θ_(h)[k−1]+mΔθ,ε_(h)[k−1]+lΔε), where m=−1,0,1 and l=1,0,1.

In step 930, for each rotation defined by the candidate anglecombination (θ_(v) ^(m), ε_(v) ^(l), θ_(h) ^(m), ε_(h) ^(l)) calculatedin step 920, a pre-selected metric is computed. The metric may be thesignal-to-noise ratio (SNR), envelope stability (ENV), differentialpower (ΔP), or other metrics known to one of skill in the signalprocessing art. The selection of the particular metric may be matched tothe coding format of the signal. n a preferred embodiment, the ENVmetric is selected to optimize the transformed signal. The ENV metric iscalculated for each polarization as

$\begin{matrix}{\begin{pmatrix}{ENV}_{v}^{ml} \\{ENV}_{h}^{ml}\end{pmatrix} = \begin{pmatrix}{\mu_{V}^{2}/\sigma_{V}^{2}} \\{\mu_{H}^{2}/\sigma_{H}^{2}}\end{pmatrix}} & (18) \\{{{where}\mspace{14mu}\mu_{V}} = {\frac{1}{P}{\sum\limits_{p = 0}^{P - 1}{{\left( {A^{ml}\mspace{14mu} B^{ml}} \right) \cdot \begin{pmatrix}{v^{\prime}\lbrack p\rbrack} \\{h^{\prime}\lbrack p\rbrack}\end{pmatrix}}}^{2}}}} & (19) \\{\sigma_{V}^{2} = {\frac{1}{P}{\sum\limits_{p = 0}^{P - 1}\left( {{{\left( {A^{ml}\mspace{14mu} B^{ml}} \right) \cdot \begin{pmatrix}{v^{\prime}\lbrack p\rbrack} \\{h^{\prime}\lbrack p\rbrack}\end{pmatrix}}}^{2} - \mu_{V}} \right)^{2}}}} & (20) \\{\mu_{H} = {\frac{1}{P}{\sum\limits_{p = 0}^{P - 1}{{\left( {C^{ml}\mspace{14mu} D^{ml}} \right) \cdot \begin{pmatrix}{v^{\prime}\lbrack p\rbrack} \\{h^{\prime}\lbrack p\rbrack}\end{pmatrix}}}^{2}}}} & (21) \\{\sigma_{H}^{2} = {\frac{1}{P}{\sum\limits_{p = 0}^{P - 1}\left( {{{\left( {C^{ml}\mspace{14mu} D^{ml}} \right) \cdot \begin{pmatrix}{v^{\prime}\lbrack p\rbrack} \\{h^{\prime}\lbrack p\rbrack}\end{pmatrix}}}^{2} - \mu_{H}} \right)^{2}}}} & (22)\end{matrix}$The coefficients A^(ml), B^(ml), C^(ml), and D^(ml) are defined inrelevance with equations 10–13 asA^(ml)=T₁₁, B^(ml)=T₁₂, for θ_(v)=θ_(v) ^(m) and ε_(v)=ε_(v) ^(l)C^(ml)=T₂₁, D^(ml)=T₂₂, for θ_(h)=θ_(h) ^(m) and ε_(h)=ε_(h) ^(l)The components of ENV given in equations 19–22 are averages over a totalof P symbols, the first of a continuum of P, preferably consecutivesymbols, being arbitrarily assigned an index of p=0 for these equations.The selection of the total number P of such symbols being used tocalculate these metrics depends on system requirements and otherparameters, as is known to those skilled in the art.

In step 940, the resulting values ENV_(v) ^(ml) and ENV_(h) ^(ml) andtheir associated angle pairs (θ_(v) ^(m), ε_(v) ^(l)) and (θ_(h) ^(m),ε_(h) ^(l)) are stored.

In step 950, a determination is made to see whether all possiblecandidate search steps have been evaluated. If not, steps 920–950 arerepeated for another one of the candidate search steps. If, at step 950,it is determined that all nine possible candidate search steps have beenevaluated, control goes to step 960.

In step 960, the optimum metric(s) are identified, and the two optimalrotation angle pairs (θ_(v) ^(o), ε_(v) ^(o)) and (θ_(h) ^(o), ε_(h)^(o)) are selected according to an optimization criterion. In thepreferred embodiment, the optimization criterion is the maximum ENVvalue. The rotation angles for this iteration θ_(v)[k], ε_(v)[k],θ_(h)[k], and ε_(h)[k] are also update with the optimal angles θ_(v)^(o), ε_(v) ^(o), θ_(h) ^(o), and θ_(h) ^(o) that maximize ENV_(v) andENV_(h) values. And these are used to update the T matrix usingEquations 10–13 given above.

In step 970, a determination is made to see whether a loop terminatingcondition has been met. The terminating condition may be reached whenthere are no more changes in the optimal angles after an iteration, orit may be reached upon executing a predetermined number of iterations.Other terminating conditions may also be employed. If the terminatingcondition is not met, the process continues with the next iteration. If,on the other hand, the terminating condition is met, then control flowsto step 980.

In step 980, the updated, optimum rotation angles are used to create thecoefficients for the T matrix. For the embodiment corresponding to FIGS.5 a & 5 c, this means that the coefficients of the rotation matrix T[k]are computed using the optimal angles θ_(v) ^(o), ε_(v) ^(o), θ_(h)^(o), and θ_(h) ^(o). The entire process 900 is repeated at apredetermined rate so long as that subchannel receiver is in operation.

In the above description of the flow diagram 900, the search step sizecalculated in step 920 was kept constant. In an alternate embodiment,the search step size may be adaptively adjusted with every searchiteration, the size depending on one of several parameters, such as thecalculated metric. When the calculated metric is the ENV metricdiscussed above, a step size Δθ=Δε=Z[k] can be used for the kth searchiteration, with the value Z[k], given in degrees, being given by

$\begin{matrix}{{Z\lbrack k\rbrack} = \left\{ \begin{matrix}{1,{{{if}\mspace{14mu}{{ENV}\left\lbrack {k - 1} \right\rbrack}} > {ENV}_{target}}} \\\left\lfloor {{\left. {3\left( {{ENV}_{target} - {{ENV}\left\lbrack {k - 1} \right\rbrack}} \right.} \right\rfloor + 1},\mspace{14mu}{otherwise}} \right.\end{matrix} \right.} & (23)\end{matrix}$where ENV[k−1] is the maximum ENV from the previous search iteration andENV_(target) is a preset target value. The selection of the target valuemay be based on the desired bit error rate for the system. In thisexample, Z preferably is constrained to a predetermined range, such asthe range [0:30].Polarization Controller PMD Compensation Algorithm

FIG. 9 b is a flow diagram 902 of the PMD compensation control algorithmwithin the PMD compensation control module 596 associated with receiverprocessor 575, for the embodiment corresponding to FIGS. 5 b & 5 d. ThePMD compensation control module 596 operates on the compensated digitalsignals 454 a, 454 b, representing h[p] & v[p] rather than on theuncompensated digital values h′[p], v′[p]. In this embodiment, theoptimal angles are output to the PMD compensator 538 to effect thenecessary correction.

The PMD device control signals 585 from the PMD compensation controlmodule 596 associated with processor 575 (FIGS. 5 b & 5 d) include apair signals designated for present purposes as Cv1 and Cv2. The currentstate of the PMD compensator is defined by the control signals Cv1 andCv2, where Cv1 is proportional to θ/2, and Cv2 is proportional to ε/2.The control algorithm peforms an iterative search routine to find theoptimum angles.

Each iteration involves testing 9 possible SOP rotations (as in thedigital case). The ENV metric is computed for each test and its value isstored along with the rotation angles. The iteration is completed whenall 9 tests have been completed. Then, the optimal ENV metric isselected and the control signals to the PMD compensator are updated withthe optimal ones. Then next iteration begins. The control signals to thePMD compensator 538 are updated for each test step in each iteration.This differs from the “all-digital” case where the final updates aremade only when search algorithm has converged.

After commencing in step 914, in step 924, the algorithm computes acandidate step size Z(k), preferably dependent on the loop iterationnumber k.

In step 925, the next iteration step, i.e., the next pair of the ninecandidate angles is selected.

In step 935, the ENV metric is computed by the PMD compensation controlmodule 596 using signals h[p], v[p] from lines 454 a, 454 b,respectively, as follows:

${{ENV}_{H}^{ml} = \frac{\mu_{H}^{2}}{\sigma_{H}^{2}}},{{ENV}_{V}^{ml} = \frac{\mu_{V}^{2}}{\sigma_{V}^{2}}},{where}$${\mu_{H} = {\frac{1}{P}{\sum\limits_{p = 0}^{P - 1}{{h\lbrack p\rbrack}}^{2}}}},{\sigma_{H}^{2} = {\frac{1}{P}{\sum\limits_{p = 0}^{P - 1}\left( {{{h\lbrack p\rbrack}}^{2} - \mu_{H}} \right)^{2}}}},{and}$$\;{{\mu_{V} = {\frac{1}{P}{\sum\limits_{p = 0}^{P - 1}{{v\lbrack p\rbrack}}^{2}}}},{\sigma_{V}^{2} = {\frac{1}{P}{\sum\limits_{p = 0}^{P - 1}{\left( {{{v\lbrack p\rbrack}}^{2} - \mu_{V}} \right)^{2}.}}}}}$

In step 945, the calculated metrics and associated angles are stored andthe control signals Cv1 and Cv2 are calculated from these angles.

In step 955, the processor 575 outputs the control signals Cv1 and Cv2(which, as stated above, belong to the suite of controls signalsgenerally shown as 585) to the PMD compensator 528, thereby adjustingthe instantaneous values of h[p] and v[p].

In step 965, a determination is made to see whether all nine candidateangle pairs have been tried. If not, control returns to step 925 tocompute the next angle pair. If, on the other hand, all nine pairs havebeen attempted, control goes to step 975.

In step 975, the optimum metric and associated angles are determinedfrom among the nine calculated sets, and the corresponding optimumcontrol signals Cv1 and Cv2 are calculated

Finally, in step 985, the PMD compensator 538 is updated using theoptimum control signals Cv1 and Cv2.

Again, the entire process 902 is repeated while the receiver isoperating.

Symbol Timing Recovery

The symbol time recovery module can include a Stokes-based timing errordetector module (“Stokes module”), a Mueller & Muller timing errordetector module (“M&M module”), or be a hybrid comprising both.

The Stokes module does not require PMD-compensated signals to provide asufficiently accurate signal to create clock adjustment signal for useby the optical detectors 700 a, 700 b. Instead, the Stokes module canuse the uncompensated digital signals 452 a, 452 b (h′[v] & v′[p]).Thus, the Stokes module is well-suited for the “all-digital” embodimentcorresponding to FIGS. 5 a & 5 c. The output of the Stokes module isused to generate the timing error signal, τ_(s), that is translated intoa clock adjustment signal directed towards minimization of the varianceof the computed Stokes parameters, thereby leading to a reduction of thetiming error,τ_(s).

The Stokes module within the symbol timing recovery module is based onthe estimation of the variance of Stokes parameters obtained from thedigitized signals 452 a, 452 b. In general, the variance of the Stokesparameters is minimal when the timing error is zero, and it grows as thetiming error increases. The Stokes algorithm within the symbol timingrecovery module results in the processor outputting clock adjustmentsignals 774 a, 774 b as part of the suite of control signals 581.Applying the clock adjustment signals 774 a, 774 b to the Integrate &Dump units 760 a, 760 b of FIG. 7, leads to a minimization of varianceof the Stokes parameters.

The Stokes based timing error detector computes the discriminatorfunction, DF(τ). In a preferred embodiment, the discriminator functionis given by

$\begin{matrix}{{{DF}\left( \tau_{s} \right)} = {1 - {\frac{1}{P}{\sum\limits_{p = 0}^{P - 2}{\left( {{S\lbrack p\rbrack} \circ {S\left\lbrack {p + 1} \right\rbrack}} \right)}}}}} & (24)\end{matrix}$where τ_(s) is the normalized timing error and the inner product of thetwo Stokes vectors is given byS[p]∘S[p+1]=s ₁ [p]s ₁ [p+1]+s ₂ [p]s ₂ [p+1]+s ₃ [p]s ₃ [p+1].  (25)In equation 24, P is the number of inner products of the most recentStokes vectors used to estimate the variance. In general, the value of Pis preset and is selected to satisfy the system requirements.

FIG. 10 shows the discriminator function of equation 24 where P=1280.More particularly, FIG. 10 shows a symmetric discriminator function witha minimum when the timing error is zero.

Based on the discriminator function output, the Stokes recoveryalgorithm computes the timing error signal τ_(s). The timing errorsignal τ_(s) is translated to a clock adjustment signal directed towardsthe reduction of the estimated error. The discriminator function outputdoes not provide the sign of the timing error. To mitigate the signambiguity, the symbol recovery algorithm, for example, may use agradient search method to initially find the occurrence of the minimumvariance and use a dithering technique to track that minimum as thesymbol timing may drift.

Unlike the Stokes module, the M&M module uses the polarizationcompensated signals, h[p] and v[p], rather than the uncompensatedsignals h′[p] and v′[p]. More particularly, the M&M module uses thecompensated signals to estimate a timing error, τ_(M). The timing error,τ_(M) is then translated into a clock adjustment signal 774 a, 774 bwhich is directed towards driving the timing error estimate τ_(M) tozero. Implementation of the M&M module is well-known to those of skillin the signal processing art and so is not described here in furtherdetail. However, it should be understood that, as used herein, the term“M&M module” is representative of any detector capable of recovering thesymbol timing of a PMD-compensated received signal, known to thoseskilled in the art of digital receivers.

The symbol timing recovery module may comprise a hybrid of a Stokesmodule and an M&M module. In a hybrid symbol timing recovery module,when uncompensated signals h′[p] & v′[p] are input to the processor(such as processor 560 in FIG. 5 a), the hybrid module is typicallyprogrammed or otherwise configured to first invoke the Stokes moduleduring a first period of time, and then invoke the M&M modulethereafter. The reason for this protocol is now explained.

When the subchannel receiver 510 of FIG. 5 a begins to process receivedoptical signals, the uncompensated signals h′[p] & v′[p] are madeavailable to the processor 560. As this happens, the PMD compensationmodule within the processor is invoked to calculate the various T-matrixcoefficients for use in Equations 1–3 to help produce the compensatedsignals h[p] & v[p].

However, it takes some time for the PMD compensation module tointernally output accurate values of the compensated signals h[p] &v[p]. During this start-up time in which the compensated signals h[p] &v[p] are of insufficient quality (e.g., not very accurate), the M&Mmodule cannot be used to create symbol timing recovery signals, such asthe clock adjustment signals 774 a, 774 b. Therefore, during thisstart-up time, the PMD compensation module invokes the Stokes module toprocess the uncompensated signals h′[p] & v′[p] and output theappropriate clock adjustment signals 774 a, 774 b to drive the I &Dunits 760 a, 760 b. After the start-up time, however, the values of thecompensated signals h[p] & v[p] are more accurate, and the symbol timingrecovery module within the processor has the M&M module operate on the“more accurately” compensated signals h[p] & v[p] to output the neededclock adjustment signals 774 a, 774 b.

The length of the startup time is based on the time needed to ensurethat the PMD compensation algorithm has converged to an optimumcompensation solution, after which the M&M module is used. The durationof the start-up time may be based on system parameters, or may simply beselected to be correspond to when an aforementioned terminatingcondition is reached.

In general, using the M&M module results in a lower bit-error rate (BER)because the timing error signal generated by the M&M module is moreaccurate than the one generated by the Stokes module. Therefore, it ispreferred that a hybrid symbol detection module first invoking theStokes module and later the M&M module be used in those cases where theprocessor receives uncompensated signals as input, such as discussedabove with respect to FIG. 5 a. However, in those cases where theprocessor receives compensated signals as input, such as in theembodiment seen in FIG. 5 b, the M&M module alone preferably is used.

The invention described and claimed herein is not to be limited in scopeby the preferred embodiments herein disclosed, since these embodimentsare intended as illustrations of several aspects of the invention. Anyequivalent embodiments are intended to be within the scope of thisinvention. Indeed, various modifications of the invention in addition tothose shown and described herein will become apparent to those skilledin the art from the foregoing description. Such modifications are alsointended to fall within the scope of the appended claims.

1. An optical communication receiver comprising: an optical splitterhaving a splitter input and a plurality of splitter outputs, the opticalsplitter configured to receive an optical channel signal comprising anumber K subchannel signals, and output K identical received channelsignals; K subchannel receivers, the subchannel receiver comprisingoptical and digital circuitry configured to receive the k^(th) of said Kidentical received channel signals and a reference light beam having asubchannel frequency f_(k), and output a first digital signalrepresentative of in-phase and quadrature components of a firstorthogonal polarization component associated with the subchannelfrequency f_(k), and also output a second digital signal representativeof in-phase and quadrature components of a second orthogonalpolarization component associated with the subchannel frequency f_(k),the first and second digital signals containing informationrepresentative of a data stream used to modulate the k^(th) subchannelfrequency; and a receiver processor configured to receive said first andsecond digital signals and output said data stream.
 2. The opticalcommunication receiver of claim 1, wherein the receiver furthercomprises: a frequency calibration circuit configured to calibrate atleast one light source to thereby maintain a frequency spacing of Δfbetween K adjacent subchannel light beams, the frequency calibrationcircuit receiving, as input, at least K subchannel light beams eachcharacterized by a subchannel frequency f_(k) and outputting at leastone frequency calibration control signal applied to at least one lightsource creating at least one of said K subchannel light beams.
 3. Theoptical communication receiver of claim 2, wherein the frequencycalibration circuit comprises: a first optical switch configured toselect from among (a) said K subchannel light beams and (b) a referencelight beam, to thereby output a selected beam; an optical splitterconfigured to split the selected signal to first and second selectedsplit beams; an optical delay configured to receive the first selectedsplit beam as input, delay the first selected split beam by apredetermined time delay T, and output a delayed first selected splitbeam; an optical detector configured to receive the delayed firstselected split beam and the second selected split beam, and output adigitized electrical signal that is proportional to e^(jωT), where ω isthe frequency of the selected beam; a controller configure to receivethe digitized electrical signal and output said at least one frequencycalibration control signal.
 4. The optical communication receiver ofclaim 1, wherein the k^(th) subchannel receiver comprises: apolarization beam splitter configured to receive and split the k^(th) ofsaid K identical receiver channel signals into first and secondorthogonal polarization components; a first optical phase detectorconfigured to receive the first orthogonal polarization component andthe reference light beam having a subchannel frequency f_(k) as inputs,and output said first digital signal; a second optical phase detectorconfigured to receive the second orthogonal polarization component andthe reference light beam having a subchannel frequency f_(k) as inputs,and output said second digital signal, and wherein the first and seconddigital signals are input to said receiver processor.
 5. The opticalcommunication receiver of claim 4, wherein the k^(th) subchannelreceiver further comprises: a first variable optical delay configured toselectively delay the first orthogonal polarization component before itis input to the first optical phase detector; a second variable opticaldelay configured to selectively delay the second orthogonal polarizationcomponent before it is input to the second optical phase detector,wherein the first and second variable optical delays are controlled bysaid receiver processor.
 6. The optical communication receiver of claim4, wherein each optical phase detector includes first and secondintegrate and dump filters configured to integrate detected analogrespective in-phase and quadrature signals for an integrating periodthat is less than a symbol period to thereby produce respective detectedanalog in-phase and quadrature subchannel signals corresponding to saidsubchannel frequency f_(k).
 7. The optical communication receiver ofclaim 1, wherein the receiver processor comprises: a polarization modedispersion (PMD) compensation control module configured to digitallycompensate the first and second digital signals to thereby producePMD-compensated first and second digital signals; a synchronization andsymbol timing module configured to produce optical detector controlsignals including at least one clock signal for controlling thesubchannel receiver, based on at least one of the first and seconddigital signals and the first and second PMD-compensated digitalsignals; and a data demodulation module configured to output the datastream that was used to modulate at least one of said K subchannels,based on the first and second PMD-compensated digital signals.
 8. Theoptical communication receiver of claim 7, wherein the synchronizationand symbol timing module includes a Stokes-based timing error detectormodule.
 9. The optical communication receiver of claim 7, wherein thesynchronization and symbol timing module also includes a Mueller &Muller timing error detector module, and wherein the Stokes-based timingerror detector module is invoked first and the Muller and Muller timingerror detector module is invoked thereafter.
 10. The opticalcommunication receiver of claim 7, wherein the PMD compensation controlmodule is configured to execute an iterative search procedure to findoptimum coefficients of a rotation matrix for rotating the first andsecond digital signals to thereby create the PMD-compensated first andsecond digital signals.
 11. The optical communication receiver of claim10, wherein, during each iteration, the search procedure calculates atleast one metric for each of a plurality of candidate pairs of rotationangles, and selects the candidate pair of rotation angles correspondingto an optimization criterion for said at least one metric, to therebycalculate the coefficients of said rotation matrix, the iterationscontinuing 5 until a terminating condition is met.
 12. The opticalcommunication receiver of claim 11, wherein a step size of the candidatepairs of rotation angles is adjusted at each iteration.
 13. The opticalcommunication receiver of claim 7, wherein the receiver processorfurther comprises a frequency offset compensator.
 14. The opticalcommunication receiver of claim 1, wherein the subchannel receivercomprises: a polarization mode dispersion (PMD) compensator deviceconfigured to receive the k^(th) of said K identical receiver channelsignals as input and output a PMD-compensated version of said k^(th)identical receiver channel signals, the PMD compensator device beingcontrolled by at least one PMD device control signal from the receiverprocessor; a polarization beam splitter configured to receive and splitsaid PMD compensated version of said identical receiver channel signalinto first and second orthogonal polarization components; a firstoptical phase detector configured to receive the first orthogonalpolarization component and the reference light beam having a subchannelfrequency f_(k) as inputs, and output said first digital signal; asecond optical phase detector configured to receive the secondorthogonal polarization component and the reference light beam having asubchannel frequency f_(k) as inputs, and output said second digitalsignal, and wherein the first and second digital signals are input tosaid receiver processor.
 15. The optical communication receiver of claim14, wherein the subchannel receiver further comprises: a first variableoptical delay configured to selectively delay the first orthogonalpolarization component before it is input to the first optical phasedetector; a second variable optical delay configured to selectivelydelay the second orthogonal polarization component before it is input tothe second optical phase detector, wherein the first and second variableoptical delays are controlled by said receiver processor.
 16. Theoptical communication receiver of claim 14, wherein each optical phasedetector includes first and second integrate and dump filters configuredto integrate detected analog respective in-phase and quadrature signalsfor an integrating period that is less than a symbol period to therebyproduce respective detected analog in-phase and quadrature subchannelsignals corresponding to said subchannel frequency f_(k).
 17. Theoptical communication receiver of claim 14, wherein the receiverprocessor comprises: a polarization mode dispersion (PMD) compensationcontrol module configured to produce said at least one PMD devicecontrol signal, based on the first and second digital signals; asynchronization and symbol timing module configured to produce opticaldetector control signals including at least one clock signal forcontrolling the subchannel receiver, based on said first and seconddigital signals; and a data demodulation module configured to output thedata stream that was used to modulate the k^(th) subchannel, based onthe first and second digital signals.
 18. The optical communicationreceiver of claim 17, wherein the synchronization and symbol timingmodule includes Mueller & Muller timing error detector module.
 19. Theoptical communication receiver of claim 17, wherein the PMD compensationcontrol module is configured to execute an iterative search procedure tofind optimum rotation angles for producing the PMD device controlsignals.
 20. The optical communication receiver of claim 19, wherein,during each iteration, the search procedure determines a candidate pairof rotation angles, calculates said at least one metric for saidcandidate pair of rotation angles, stores the metric, and produces saidat least one PMD device control signal that is applied to the PMDcompensator device, for each candidate pair of rotation angles.
 21. Theoptical communication receiver of claim 20, wherein a step sizegoverning selection of the candidate pairs of rotation angles isadjusted at each iteration.
 22. The optical communication receiver ofclaim 17, wherein the receiver processor further comprises a frequencyoffset compensator.
 23. A frequency calibration system for calibrating anumber K of laser light beams, each laser light beam having a frequencyf_(k), k=1, 2, 3, . . . , K, the frequency calibration systemcomprising: an optical switch system configured to select one from amongthe K laser light beams and a reference beam and output a selected beam;a splitter disposed to receive the selected beam and output firstidentical first and second selected beams; an optical detectorconfigured to receive a delayed version of the first selected beam andthe second selected beam, and output at least one electrical signalproportional to a phase difference between the two beams; a controllerconfigured to receive said at least one electrical signal and output atleast one frequency calibration control signal to control at least onelight source responsible for creating at least one of said plurality oflaser light beams.
 24. The frequency calibration system of claim 23,wherein the optical switch system comprises a K:1 switch configured toselect one from among said K light beams and a 2:1 switch configured aselect from among the reference light beam and said one from among saidK light beams to thereby output said selected beam.
 25. The frequencycalibration system of claim 23, wherein the first selected 5 beam isdelayed by one symbol period.
 26. An iterative method for compensatingfor polarization mode dispersion (PMD) in an optical signal comprising:(a) determining a candidate pair of rotation angles for adjusting astate of polarization of the optical signal; (b) calculating at leastone metric for said candidate pair of rotation angles (c) storing the atleast one metric and also outputting at least one PMD device controlsignal that is applied to a PMD compensator device into which theoptical signal is input; (d) repeating steps (a), (b) and (c) untilmetrics; for a predetermined set of candidate pairs have beencalculated; (e) finding the optimum metric and the optimum rotationangles corresponding to that metric; and (f) outputting at least one PMDdevice control signal which corresponds to the optimum angles, to saidPMD compensator device into which the optical signal is input.
 27. Themethod according to claim 26, wherein the metric is an envelopestability metric.
 28. The method according to claim 26, furthercomprising repeating steps (a)–(f) until a predetermined condition ismet, and wherein a step size for the candidate pairs of rotation anglesis adjusted at each iteration of steps (a)–(f).
 29. A method forcompensating for polarization mode dispersion (PMD) in an optical signalhaving two orthogonal polarizations, the method comprising: (a)determining a candidate pair of rotation angles for adjusting a state ofpolarization of the optical signal; (b) calculating at least one metricfor said candidate pair of rotation angles; (c) storing the metric; (d)performing steps (a), (b) and (c) until metrics for a predetermined setof candidate pairs have been calculated; (e) finding the optimum metricand the optimum rotation angles corresponding to that metric, and thenupdating a rotation matrix having coefficients derived from the optimumrotation angles; (f) digitally compensating for PMD by applying therotation matrix to digitized signals representing the information set onthe two orthogonal polarizations.
 30. The method according to claim 29,wherein the metric is an envelope stability metric.
 31. The methodaccording to claim 29, further comprising, before step (f), 20 repeatingsteps (a)–(e) until a predetermined condition is met.
 32. The methodaccording to claim 31, further comprising adjusting a step size for thecandidate pairs of rotation angles at each iteration of steps (a)–(e).